Electric power steering apparatuses using motor torque to provide a power assist to steering are widely used in automobiles so as to allow a light steering operation. Such an electric power steering system includes a geared or belted transmission mechanism transmitting the driving force of a motor through a reducer to provide the power assist to a steering shaft or a rack shaft.
FIG. 1 shows a typical structure of such an electric power steering system. A column shaft 302 of a steering wheel 301 is connected to tie rods 306 of the steering control wheels via reduction gears 303, universal joints 304a and 304b and a pinion-rack mechanism 305. The column shaft 302 is provided with a torque sensor 307 for detecting the steering torque of the steering wheel 301. A motor 308 assisting the steering force of the steering wheel 301 is connected to the column shaft 302 via the reduction gears 303.
In the electric power steering system having such a structure, the torque sensor 307 detects a steering torque generated by a driver's operating force on the steering transmitted from the steering wheel 301, and the motor 308 is driven and controlled by a current command value calculated based on a torque signal and a vehicle speed signal. This drive provides the power assist to the driver's operation of the steering wheel 1 and thereby gives the driver a light steering operation. In other words, the steering feel of the steering wheel as well as the performance of the electric power steering system significantly depend on: the current command value determined on the basis of the steering torque produced by a steering wheel operation; and the method of controlling the motor 308 based on the current command value.
Before describing the problems that must be overcome to achieve preferred embodiments of the motor control for the electric power steering system from the foregoing perspective, a description will first be given of the general motor characteristics of motors used for the electric power steering systems, and the motor torque control method.
First, the motor characteristics will now be described. The normal operating range of a motor can be defined by a torque-speed characteristic (T-n characteristic) derived from a motor output equation. The motor output equation for a three-phase brushless DC-motor (BLDC motor) can be expressed by the following Formula 1:v=EMF+R·i+L·di/dt  (Formula 1)                where “v” is the phase voltage of the motor, “i” the phase current of the motor, “EMF” the phase back EMF voltage, “R” the per-phase resistance, and “L” the per-phase inductance.        
Assuming here that saturation (i.e., PWM duty=100%) is reached, a motor-driving battery voltage Vbat will be applied to the two windings of the motor. Therefore, the above Formula 1 can be transformed into Formula 2 below:Vbat=EMFLL+2R·I  (Formula 2)                where “EMFLL” is the back EMF voltage measured between two phases, and “I” is the motor current.        
Next, using Formula 3, which is an equation for the back EMF voltage EMF, and Formula 4, which is a torque equation, the following Formula 5 can be derived from Formula 2 above:EMFLL=Ke·ω  (Formula 3)                where “Ke” is the back EMF constant and “ω” is the angular speed (rotational speed).T=Kt·I  (Formula 4)        where “Kt” is the torque constant.ω=ω0(1−(I/I0))=ω0(1−(T/T0)) [rad/s]  (Formula 5)        where “ω0=Vbat/Ke” is the no-load angular speed (one under no-torque condition), “I0=Vbat/2R” the locked rotor current (i.e., stall current) (angular speed=0), and “T0=Kt·I0” the locked rotor torque (i.e., stall torque).        
A “rad/s” to “rpm” conversion of the foregoing Formula 5 yields the following Formula 6:n=n0(1−(I/I0)=n(1−(T/T0)) [rpm]  (Formula 6)The foregoing Formula 6 expresses a linear torque-speed characteristic (T-n characteristic).
Here, the actual torque-speed characteristic (T-n characteristic) of a brushless DC-motor slightly differs from Formula 6 and can be given by the following Formula 7:n=n0−(n0−nrated)·T/Trated  (Formula 7)                where “n0” is the no-load rotational speed, “nrated” the rated rotational speed, and “Trated” the rated torque.        
The above Formulas 6 and 7 can be graphically represented as shown in FIG. 2, where point “A” represents the rated value and point “B” the no-load value. The broken line given by Formula 6 is an ideal straight line, while the actual characteristic given by Formula 7 (solid curve) differs slightly from such an identical straight line. This difference is attributable to the influence of the inductance value L of the motor. The higher the applied current is, the more the actual characteristic line deviates from the ideal straight line.
The T-n characteristic in FIG. 2 represents the performance limit of the motor. Within the range below the T-n characteristic curve in FIG. 2, the motor starts from a stopped condition and runs up to the maximum angular speed to produce the maximum torque possible without exceeding thermal and electrical limits.
In FIG. 3, a characteristic line 1 represents the T-n characteristic of a low-power motor, while a characteristic line 3 represents the T-n characteristic of a high-power motor. If a characteristic curve 2 represents a motor load characteristic of the electric power steering system, a high-power motor the exhibiting characteristic 3 would cover the entire range of the load characteristic represented by the characteristic curve 2. However, use of such a motor would involve a problem of increasing cost and size. An attempt to use a low-power motor exhibiting the characteristic 1 to cover the load characteristic represented by the characteristic curve 2 would, however, leave the high-speed rotation range of the characteristic 2 uncovered. As a method for covering the load characteristic represented by the characteristic curve 2 with a motor exhibiting the characteristic 1, a field-weakening control of a motor vector control can be used to convert the T-n characteristic of the motor exhibiting the characteristic 1 into the T-n characteristic represented by the characteristic curve 4.
There are well-known conventional methods for controlling a motor in an electric power steering system by the vector control with the field-weakening control taken into consideration. For example, Japanese Patent Application Laid-open No. 2001-18822 A discloses a method for controlling a motor in an electric power steering system by a vector control.
FIG. 4 shows a basic block diagram of the electric power steering system using the vector control method disclosed in Japanese Patent Application Laid-open No. 2001-18822 A.
A current command determination means 324 determines d-axis and q-axis current command values idref and iqref based on a torque command value Tref. Meanwhile, current detection means 341 and 342 detect motor currents ia, ib and ic of a motor 308, which are converted, by a three phase-to-two phase conversion means 343, into two axis (d-q axis) currents id and iq. Subtractors 325 and 326 respectively calculate current deviations between the d-axis current command value idref and a feedback current value id and a current deviation between the q-axis current command value iqref and another feedback current value iq. The current deviations are inputted to a PI-control means 328, which in turn determines voltage command values vd and vq for reducing the current deviations to zero. The motor 308 is a three-phase motor, and the voltage command values vd and vq are converted by a two phase-to-three phase conversion means 336 into three phase voltage command values va, vb and vc.
A PWM-control means 337 generates PWM-controlled gate signals based on the three-phase voltage command values va, vb and vc. An inverter 338 is driven by the gate signals generated by the PWM-control means 337 to supply the motor 308 with currents that reduce the current deviations to zero. A resolver 316 detects and feeds the angle (rotational position) θ of the motor 308 to an angular speed conversion means 348 to determine an angular speed (rotational speed) ω used for the vector control.
In such vector control, the field-weakening control is used in the high-speed range of the motor 308.
Here, the motor 308 is controlled by the vector control on the basis of a steering assist current command value Iref determined on the basis of steering torque (or vehicle speed and so on) detected by the torque sensor 307. This vector control is given either by the following Formula 8, if no field-weakening control is performed (Id=0), or the following Formula 9, if the field-weakening control is performed (Id≠0).Iq=IrefId=0  (Formula 8)Iq=IrefId≠0  (Formula 9)
Meanwhile, a motor current Is is given in terms of a d-axis current command value Id and a q-axis current command value Iq by Formula 10 below:Is=√(Iq2+Id2)  (Formula 10)
An abrupt turning-returning operation of the steering wheel that provides the vector control over a motor having such a current relationship as a condition will cause the motor to fail to produce the necessary torque. Consequently, the motor will remain in a range with the field-weakening control. In other words, a PWM-duty saturation (duty=100%) may occur in the high-speed rotation range.
The PWM-duty saturation results in a current waveform distortion, causing a large torque ripple in the motor, as a result of which, steering wheel vibration or abnormal motor noise is caused.
Such an attempt to perform the control beyond the limit of a rated torque output will result in the PWM-duty saturation and consequently a large torque ripple, causing the driver to experience steering wheel vibration or an unnatural steering feel.
To deal with such defective conditions, control methods have been proposed including one according to Japanese Patent Application Laid-open No. 08-142886 A. FIG. 5 shows a control block diagram disclosed in Japanese Patent Application Laid-open No. 08-142886 A.
When a motor driving signal SM calculated on the basis of a motor current command value SI calculated by a current command calculator 2018 is between an upper limit value SMAX and a lower limit value −SMAX, the motor 308 will be PWM-controlled according to the calculated motor driving signal SM. However, when the motor driving signal SM calculated on the basis of the motor current command value SI is equal to or over the upper limit value SMAX or is equal to or below the lower limit value −SMAX, the motor driving signal SM will be replaced with the upper limit value SMAX or the lower limit value −SMAX so that the output value will be limited by a limiter 2110.
Thus, in the apparatus according to Japanese Patent Application Laid-open No. 08-142886 A, the PWM-duty is forcibly limited to prevent the PWM-duty saturation.
However, in the apparatus according to Japanese Patent Application Laid-open No. 08-142886 A, an output value to a PWM-circuit 2029 is limited by the limiter 2110, and hence the driver will still be left with an unnatural steering feel.
Then, there are three concepts to be understood with regard to the field-weakening control for the permanent magnet type brushless DC-motors. These concepts are a constant power control with a constant voltage, a constant power control with a constant current and a constant voltage control with a constant current. The first two concepts are relatively simple, but the constant power conditions cannot be maintained over a wide range of speed. The third concept takes the limit conditions into consideration and hence is more accurate. However, motor resistance is often ignored, resulting in significant calculation errors with, in particular, small motors. The motor resistance is relatively high with respect to inductance. That is, the influence of not only the motor resistance but also the inductance must be taken into consideration.
In the electric power steering system, the steering assist torque is a function of steering torque. When the steering wheel is turned back very quickly, a required steering assist torque will be too large for the motor to follow immediately. Therefore, the motor cannot provide an accurate steering assist torque. This means that a motor current limiting system such as the one described above (involving the limited motor power and the duty saturation) cannot provide a motor current corresponding to the current command value (reference current) directly proportional to the required torque. In this condition, a torque ripple occurs resulting in an unnatural steering feel or abnormal noise.
The present invention has been made in view of the aforementioned circumstances. An object of the present invention is to provide a method and apparatus for controlling an electric power steering system in which a current command value is limited to set a d-axis current as low as possible within a range meeting the requirement specifications at a time of a d-axis field-weakening control of a vector control so that a torque ripple in a high-speed range of the motor during an abrupt turning-returning operation of the steering wheel can be reduced, thereby preventing steering wheel vibration and an uncomfortable steering feel. This is because there is a problem of excessive noise caused by such factors as torque ripple due to overuse of the field-weakening control beyond the performance limits of the motor.
Further, an object of the present invention is to provide a method for limiting a reference current that minimizes a torque ripple during a motor operation, and to calculate a d-axis current in the field-weakening control range based on the concepts of using an inverter and maximum functionalization to expand the operating range of the motor in order to provide a method and apparatus for controlling the electric power steering system having a very small torque ripple and a natural steering feel.